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LTC3557/LTC3557-1 USB Power Manager with Li-Ion Charger and Three Step-Down Regulators FEATURES
DESCRIPTION
The LTC(R)3557/LTC3557-1 is a highly integrated power management and battery charger IC for single cell Li-Ion/ Polymer battery applications. It includes a PowerPathTM manager with automatic load prioritization, a battery charger, an ideal diode and numerous internal protection features. Designed specifically for USB applications, the LTC3557/LTC3557-1 power manager automatically limits input current to a maximum of either 100mA or 500mA for USB applications or 1A for wall adapter powered applications. Battery charge current is automatically reduced such that the sum of the load current and the charge current does not exceed the programmed input current limit. The LTC3557/LTC3557-1 also includes three adjustable synchronous step-down switching regulators and a high voltage buck regulator output controller with Bat-Track that allows efficient charging from supplies as high as 38V. The LTC3557/LTC3557-1 is available in a low profile 4mm x 4mm x 0.75mm 28-pin QFN package.
, LT LTC, LTM and Burst Mode are registered trademarks of Linear Technology , Corporation. Bat-Track and PowerPath are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 6522118, 6700364. Other patents pending.

Seamless Transition Between Input Power Sources: Li-Ion Battery, USB, 5V Wall Adapter or High Voltage Buck Regulator with Bat-TrackTM 200m Internal Ideal Diode Plus Optional External Ideal Diode Controller Provides Low Loss Power Path When Input Current is Limited or Unavailable Triple Adjustable High Efficiency Step-Down Switching Regulators (600mA, 400mA, 400mA IOUT) Pin Selectable Burst Mode(R) Operation Full Featured Li-Ion/Polymer Battery Charger 1.5A Maximum Charge Current with Thermal Limiting Battery Float Voltage: 4.2V (LTC3557) 4.1V (LTC3557-1) Low Profile 4mm x 4mm 28-Pin QFN Package
APPLICATIONS

HDD-Based MP3 Players PDA, PMP PND/GPS , USB-Based Handheld Products
TYPICAL APPLICATION
HV SUPPLY 8V TO 38V (TRANSIENTS TO 60V)
Input and Battery Current vs Load Current
HIGH VOLTAGE BUCK DC/DC 600 500 400 0V CURRENT (mA) VOUT CC/CV CHARGER LTC3557/LTC3557-1 ALWAYS ON LDO TRIPLE HIGH EFFICIENCY STEP-DOWN SWITCHING REGULATORS
35571 TA01a
RPROG = 2k RCLPROG = 2k
IIN
100mA/500mA 1000mA USB OR 5V ADAPTER
ILOAD 300 200 100 0 -100 0 WALL = 0V 100 200 IBAT (DISCHARGING) 400 300 ILOAD (mA) 500 600 IBAT (CHARGING)
CHARGE
+
NTC
SINGLE CELL Li-Ion 3.3V/25mA 0.8V to 3.6V/600mA 0.8V to 3.6V/400mA 0.8V to 3.6V/400mA
RST
35571 TA01b
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LTC3557/LTC3557-1 ABSOLUTE MAXIMUM RATINGS
(Notes 1-4)
PIN CONFIGURATION
CLPROG TOP VIEW ACPR VBUS VOUT FB2 BAT 21 GATE 20 PROG 19 NTC 29 18 VNTC 17 SW3 16 VIN2 15 SW2 8 MODE 9 10 11 12 13 14 EN1 EN2 EN3 FB3 RST2 VC CHRG ILIM0 1 ILIM1 2 WALL 3 LDO3V3 4 SW1 5 VIN1 6 FB1 7
VBUS, VOUT, VIN1, VIN2 t < 1ms and Duty Cycle < 1% .................. -0.3V to 7V Steady State ............................................. -0.3V to 6V BAT, NTC, CHRG, WALL, VC, MODE, FB1, FB2, FB3, RST2 ........................ -0.3V to 6V EN1, EN2, EN3 .............................. -0.3V to VOUT + 0.3V ILIM0, ILIM1, PROG ....................... -0.3V to VCC + 0.3V IVBUS, IVOUT, IBAT .........................................................2A ISW1 .....................................................................850mA ISW2, ISW3 ............................................................600mA IRST2, ICHRG, IACPR .................................................75mA ICLPROG, IPROG..........................................................2mA Maximum Operating Junction Temperature .......... 110C Operating Ambient Temperature Range ... -40C to 85C Storage Temperature Range................... -65C to 125C
28 27 26 25 24 23 22
UF PACKAGE 28-LEAD (4mm 4mm) PLASTIC QFN TJMAX = 110C, JA = 37C/W EXPOSED PAD (PIN 29) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3557EUF#PBF LTC3557EUF-1#PBF TAPE AND REEL LTC3557EUF#TRPBF LTC3557EUF-1#TRPBF PART MARKING 3557 35571 PACKAGE DESCRIPTION 28-Lead (4mm x 4mm) Plastic QFN 28-Lead (4mm x 4mm) Plastic QFN TEMPERATURE RANGE -40C to 85C -40C to 85C
Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
POWER MANAGER ELECTRICAL CHARACTERISTICS
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VBUS = 5V, VBAT = 3.8V, ILIM0 = ILIM1 = 5V, WALL = EN1 = EN2 = EN3 = 0V, RPROG = 2k, RCLPROG = 2.1k.
SYMBOL VBUS IBUS(LIM) PARAMETER Input Supply Voltage Total Input Current (Note 5) ILIM0 = 0V, ILIM1 = 0V (1x Mode) ILIM0 = 5V, ILIM1 = 5V (5x Mode) ILIM0 = 5V, ILIM1 = 0V (10x Mode) 1x, 5x, 10x Modes ILIM0 = 0V, ILIM1 = 5V (Suspend Mode)

CONDITIONS
MIN 4.35 80 450 900
TYP
MAX 5.5
UNITS V mA mA mA mA mA mA/mA V V V
Input Power Supply 90 475 950 0.35 0.05 1000 0.2 1.0 2.0 3.5 3.8 3.7 3.9 100 500 1000 0.1
IBUSQ hCLPROG VCLPROG
Input Quiescent Current
Ratio of Measured VBUS Current to CLPROG 1x, 5x, 10x Modes Program Current CLPROG Servo Voltage in Current Limit 1x Mode 5x Mode 10x Mode Rising Threshold Falling Threshold
VUVLO
VBUS Undervoltage Lockout
V V
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LTC3557/LTC3557-1 POWER MANAGER ELECTRICAL CHARACTERISTICS
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VBUS = 5V, VBAT = 3.8V, ILIM0 = ILIM1 = 5V, WALL = EN1 = EN2 = EN3 = 0V, RPROG = 2k, RCLPROG = 2.1k.
SYMBOL VDUVLO RON_LIM PARAMETER VBUS to VOUT Differential Undervoltage Lockout Input Current Limit Power FET On-Resistance (Between VBUS and VOUT) VBAT Regulated Output Voltage LTC3557 LTC3557, 0C TA 85C LTC3557-1 LTC3557-1, 0C TA 85C RPROG = 1k, Input Current Limit = 2A RPROG = 2k, Input Current Limit = 1A RPROG = 5k, Input Current Limit = 400mA VBUS > VUVLO, Charger Off, IOUT = 0A VBUS = 0V, IOUT = 0A (Ideal Diode Mode) BAT < VTRKL BAT < VTRKL BAT Rising BAT Falling Threshold Voltage Relative to VFLOAT Timer Starts when BAT = VFLOAT - 50mV BAT < VTRKL (Note 6) 40 2.5 -75 3.2 0.4 0.085

CONDITIONS Rising Threshold Falling Threshold
MIN
TYP 50 -50 0.2
MAX 100
UNITS mV mV
Battery Charger VFLOAT 4.179 4.165 4.079 4.065 950 465 180 4.200 4.200 4.100 4.100 1000 500 200 6 55 1.000 0.100 1000 50 2.85 2.75 -100 4 0.5 0.1 200 110 60 3.0 -115 4.8 0.6 0.115 4.221 4.235 4.121 4.135 1050 535 220 27 100 V V V V mA mA mA A A V V mA/mA mA V V mV Hour Hour mA/mA m C
ICHG
Constant Current Mode Charge Current
IBAT
Battery Drain Current
PROG Pin Servo Voltage VPROG VPROG(TRKL) PROG Pin Servo Voltage in Trickle Charge hPROG ITRKL VTRKL VRECHRG tTERM tBADBAT hC/10 RON(CHG) TLIM NTC VCOLD VHOT VDIS INTC Ideal Diode VFWD RDROPOUT IMAX VLDO3V3 RCL(LDO3V3) Forward Voltage Detection Diode On-Resistance, Dropout Diode Current Limit Regulated Output Voltage Closed-Loop Output Resistance Cold Temperature Fault Threshold Voltage Hot Temperature Fault Threshold Voltage NTC Disable Threshold Voltage NTC Leakage Current Ratio of IBAT to PROG Pin Current Trickle Charge Current Trickle Charge Rising Threshold Trickle Charge Falling Threshold Recharge Battery Threshold Voltage Safety Timer Termination Period Bad Battery Termination Time End-of-Charge Indication Current Ratio Battery Charger Power FET On-Resistance (Between VOUT and BAT) Junction Temperature in Constant Temperature Mode
Rising NTC Voltage Hysteresis Falling NTC Voltage Hysteresis Falling NTC Voltage Hysteresis NTC = VBUS = 5V
75 34 1.2 -50
76 1.3 35 1.3 1.7 50
77 36 2.2 50
%VVNTC %VVNTC %VVNTC %VVNTC %VVNTC mV nA
IOUT = 10mA IOUT = 1A (Note 7) 0mA < ILDO3V3 < 25mA BAT = 3.0V, VBUS = 0V
5
15 200 3.6
25
mV m A
Always On 3.3V Supply 3.1 3.3 24 3.2 3.5 V
35571fc
ROL(LDO3V3) Open-Loop Output Resistance
3
LTC3557/LTC3557-1 POWER MANAGER ELECTRICAL CHARACTERISTICS
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VBUS = 5V, VBAT = 3.8V, ILIM0 = ILIM1 = 5V, WALL = EN1 = EN2 = EN3 = 0V, RPROG = 2k, RCLPROG = 2.1k.
SYMBOL Wall Adapter VACPR VW VW IQWALL ACPR Pin Output High Voltage ACPR Pin Output Low Voltage Absolute Wall Input Threshold Voltage Differential Wall Input Threshold Voltage Wall Operating Quiescent Current IACPR = 1mA IACPR = 1mA WALL Rising WALL Falling WALL - BAT Falling WALL - BAT Rising IWALL + IVOUT, IBAT = 0mA, WALL = VOUT = 5V ILIM0, ILIM1 ILIM0, ILIM1 ILIM0, ILIM1; VPIN = 1V ICHRG = 10mA BAT = 4.5V, CHRG = 5V 1.2 2 0.15 0 0.4 1 VOUT - 0.3 VOUT 0 4.3 3.2 25 75 440 0.3 4.45 V V V V mV mV A PARAMETER CONDITIONS MIN TYP MAX UNITS
3.1 0
150
Logic (ILIM0, ILIM1 and CHRG) VIL VIH IPD VCHRG ICHRG Input Low Voltage Input High Voltage Static Pull-Down Current CHRG Pin Output Low Voltage CHRG Pin Input Current 0.4 V V A V A
SWITCHING REGULATOR ELECTRICAL CHARACTERISTICS
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VOUT = VIN1 = VIN2 = 3.8V, MODE = EN1 = EN2 = EN3 = 0V.
SYMBOL VIN1, VIN2 VOUT UVL0 PARAMETER Input Supply Voltage VOUT Falling VOUT Rising Oscillator Frequency Input Low Voltage Input High Voltage Static Pull-Down Current Pulse-Skip Mode Input Current (Note 10) Burst Mode Input Current (Note 10) Shutdown Input Current ILIM1 VFB1 IFB1 D1 RP1 RN1 RSW1(PD) Peak PMOS Current Limit Feedback Voltage FB1 Input Current (Note 10) Maximum Duty Cycle RDS(ON) of PMOS RDS(ON) of NMOS SW1 Pull-Down in Shutdown MODE, EN1, EN2, EN3 MODE, EN1, EN2, EN3 MODE, EN1, EN2, EN3 (VPIN = 1V) IOUT = 0, EN1 = 3.8V, MODE = 0V IOUT = 0, EN1 = MODE = 3.8V IOUT = 0, EN1 = 0V, FB1 = 0V EN1 = 3.8V, MODE = 0V or 3.8V (Note 7) EN1 = 3.8V, MODE = 0V EN1 = MODE = 3.8V EN1 = 3.8V FB1 = 0V, EN1 = 3.8V EN1 = 3.8V EN1 = 3.8V EN1 = 0V

CONDITIONS (Note 9) VIN1 and VIN2 Connected to VOUT Through Low Impedance. Switching Regulators are Disabled Below VOUT UVLO
MIN 2.7 2.5
TYP
MAX 5.5
UNITS V V V MHz V V A A
Step-Down Switching Regulators 1, 2 and 3 2.7 2.8 2.25
2.9 2.59 0.4
fOSC VIL VIH IPD IVIN1
1.91 1.2
1 220 35 0.01 900 0.78 0.78 -0.05 100 0.3 0.4 10 1200 0.8 0.8 50 1 1500 0.82 0.824 0.05
Step-Down Switching Regulator 1 A A mA V V A % k
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LTC3557/LTC3557-1 SWITCHING REGULATOR ELECTRICAL CHARACTERISTICS
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VOUT = VIN1 = VIN2 = 3.8V, MODE = EN1 = EN2 = EN3 = 0V.
SYMBOL IVIN2 PARAMETER Pulse-Skip Mode Input Current (Note 10) Burst Mode Input Current (Note 10) Shutdown Input Current ILIM2 VFB2 IFB2 D2 RP2 RN2 RSW2(PD) VRST2 IRST2 VTH(RST2) tRST2 IVIN2 Peak PMOS Current Limit Feedback Voltage FB2 Input Current (Note 10) Maximum Duty Cycle RDS(ON) of PMOS RDS(0N) of NMOS SW2 Pull-Down in Shutdown Power-On RST2 Pin Output Low Voltage Power-On RST2 Pin Threshold Power-On RST2 Pin Delay Pulse-Skip Mode Input Current (Note 10) Burst Mode Input Current (Note 10) Shutdown Input current ILIM3 VFB3 IFB3 D3 RP3 RN3 RSW3(PD) Peak PMOS Current Limt Feedback Voltage FB3 Input Current (Note 10) Maximum Duty Cycle RDS(ON) of PMOS RDS(ON) of NMOS SW3 Pull-Down in Shutdown CONDITIONS IOUT = 0, EN2 = 3.8V, MODE = 0V IOUT = 0, EN2 = MODE = 3.8V IOUT = 0, EN2 = 0V, FB2 = 0V EN2 = 3.8V, MODE = 0V or 3.8V (Note 7) EN2 = 3.8V, MODE = 0V EN2 = MODE = 3.8V EN2 = 3.8V FB2 = 0V, EN2 = 3.8V EN2 = 3.8V EN2 = 3.8V EN2 = 0V IRST2 = 1mA, FB2 = 0V, EN2 = 3.8V (Note 8) From RST2 Threshold to RST2 Hi-Z IOUT = 0, EN3 = 3.8V, MODE = 0V IOUT = 0, EN3 = MODE = 3.8V IOUT = 0, EN3 = 0V, FB3 = 0V EN3 = 3.8V, MODE = 0V or 3.8V (Note 7) EN3 = 3.8V, MODE = 0V EN3 = MODE = 3.8V EN3 = 3.8V FB3 = 0V, EN3 = 3.8V EN3 = 3.8V EN3 = 3.8V EN3 = 0V

MIN
TYP 220 35 0.01
MAX
UNITS A
Step-Down Switching Regulator 2 50 1 1000 0.82 0.824 0.05 0.6 0.6 10 0.1 -8 230 220 35 0.01 600 0.78 0.78 -0.05 100 0.6 0.6 10 800 0.8 0.8 50 1 1000 0.82 0.824 0.05 0.35 1 A A mA V V A % k V A % ms A A A mA V V A % k
600 0.78 0.78 -0.05 100
800 0.8 0.8
Power-On RST2 Pin Input Current (Note 10) VRST2 = 5.5V, EN2 = 3.8V
Step-Down Switching Regulator 3
Note 1. Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2. The LTC3557/LTC3557-1 is guaranteed to meet performance specifications from 0C to 85C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3. This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperatures will exceed 110C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may result in device degradation or failure. Note 4. VCC is the greater of VBUS, VOUT or BAT.
Note 5. Total input current is the sum of quiescent current, IBUSQ, and measured current given by VCLPROG/RCLPROG * (hCLPROG + 1) Note 6. hC/10 is expressed as a fraction of measured full charge current with indicated PROG resistor. Note 7. The current limit features of this part are intended to protect the IC from short term or intermittent fault conditions. Continuous operation above the maximum specified pin current rating may result in device degradation or failure. Note 8. RST2 threshold is expressed as a percentage difference from the FB2 regulation voltage. The threshold is measured for FB2 rising. Note 9. VOUT not in UVLO. Note 10. FB high, not switching.
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LTC3557/LTC3557-1 TYPICAL PERFORMANCE CHARACTERISTICS
Input Supply Current vs Temperature
0.8 0.7 0.6 IVBUS (mA) 0.5 0.4 0.3 0.2 0.1 0 -50 -25 0 75 50 25 TEMPERATURE (C) 100 125 0 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125 0.10 VBUS = 5V 1x MODE 0.08
TA = 25C unless otherwise specified Battery Drain Current vs Temperature
0.20 0.18 0.16 0.14 3 BUCKS ENABLED 2 BUCKS ENABLED
Input Supply Current vs Temperature (Suspend Mode)
VBUS = 5V
IVBUS (mA)
IBAT (mA)
0.06
0.12 0.10 0.08 0.06 1 BUCK ENABLED N0 BUCKS ENABLED VBAT = 3.8V MODE = 3.8V -25 50 25 0 75 TEMPERATURE (C) 100 125
0.04
0.02
0.04 0.02
0 -50
35571 G01
35571 G02
35571 G03
Input Current Limit vs Temperature
1200 1100 1000 900 800 IVBUS (mA) RON (m) 700 600 500 400 300 200 100 0 -50 -25 1x MODE 50 25 0 75 TEMPERATURE (C) 100 125 5x MODE VBUS = 5V RCLPROG = 2.1k 300 10x MODE 280 260 240 220 180 160 140 120 100
Input RON vs Temperature
IOUT = 400mA 600 500 VBUS = 4.5V IBAT (mA) VBUS = 5V VBUS = 5.5V 400 300 200 100
Charge Current vs Temperature (Thermal Regulation)
0 -50
VBUS = 5V 10x MODE RPROG = 2k 50 25 75 0 TEMPERATURE (C) 100 125
-25
50 25 0 75 TEMPERATURE (C)
100
125
0 -50 -25
35571 G04
35571 G05
35571 G06
Battery Current and Voltage vs Time (LTC3557)
600 500 400 IBAT (mA) 300 200 CHRG VBAT 4 3 SAFETY TIMER 2 TERMINATION C/10 IBAT 3 4 TIME (HOUR) 5 6
35571 G07
VFLOAT Load Regulation
6 5 VBAT AND VCHRG (V) 4.22 4.20 4.18 VFLOAT (V) VFLOAT (V) 4.16 4.14 4.12 4.10 4.08 4.06 0 200 400 600 IBAT (mA) 800 1000
35571 G08
Battery Regulation (Float) Voltage vs Temperature
4.22 VBUS = 5V 10x MODE IBAT = 2mA LTC3557 4.20 4.18 4.16 4.14 4.12 LTC3557-1 4.10 4.08 4.06 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125
LTC3557
1450mAhr CELL 100 VBUS = 5V RPROG = 2k RCLPROG = 2k 0 0 2 1
LTC3557-1
1 0
35571 G09
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LTC3557/LTC3557-1 TYPICAL PERFORMANCE CHARACTERISTICS
IBAT vs VBAT
600 500 400 IBAT (mA) VFWD (V) 0.15 300 200 100 0 VBUS = 5V 10x MODE RPROG = 2k RCLPROG = 2k 2.0 2.4 2.8 3.2 3.6 VBAT (V) 4.0 4.4 0.05 5 0 0 0 0.2 0.4 0.6 IBAT (A) 0.8 1.0 1.2 0 0.2 0.4 0.6 IBAT (A) 0.8 1.0
35571 G12
TA = 25C unless otherwise specified Forward Voltage vs Ideal Diode Current (with Si2333DS External FET)
40 35 30 VFWD (mV) VBAT = 3.8V VBUS = 0V TA = 25C
Forward Voltage vs Ideal Diode Current (No External FET)
0.25 VBUS = 0V TA = 25C VBAT = 3.2V VBAT = 3.6V VBAT = 4.2V
0.20
25 20 15 10
0.10
35571 G10
35571 G11
Input Connect Waveform
VBUS 5V/DIV VOUT 5V/DIV IBUS 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.75V IOUT = 100mA RCLPROG = 2k RPROG = 2k 1ms/DIV
35571 G25
Input Disconnect Waveform
VBUS 5V/DIV VOUT 5V/DIV IBUS 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.75V IOUT = 100mA RCLPROG = 2k RPROG = 2k 1ms/DIV
35571 G26
Switching from 1x to 5x Mode
ILIM0/ILIM1 5V/DIV IBUS 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.75V IOUT = 50mA RCLPROG = 2k RPROG = 2k 1ms/DIV
35571 G27
Switching from Suspend Mode to 5x Mode
ILIM0 5V/DIV VOUT 5V/DIV IBUS 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.75V IOUT = 100mA RCLPROG = 2k RPROG = 2k ILIM1 = 5V 100s/DIV
35571 G28
WALL Connect Waveform
WALL 5V/DIV VOUT 5V/DIV IWALL 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.75V IOUT = 100mA RPROG = 2k 1ms/DIV
35571 G29
WALL Disconnect Waveform
WALL 5V/DIV VOUT 5V/DIV IWALL 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.75V IOUT = 100mA RPROG = 2k 1ms/DIV
35571 G30
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LTC3557/LTC3557-1 TYPICAL PERFORMANCE CHARACTERISTICS
Oscillator Frequency vs Temperature
2.5 2.4 2.3 2.2 fOSC (MHz) 2.1 2.0 1.9 1.8 1.7 1.6 1.5 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125 VIN = 2.9V VIN = 3.8V EFFICIENCY (%) VIN = 5V 100 Burst Mode 90 OPERATION 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 1 10 IOUT1 (mA) VOUT1 = 3.3V VIN1 = 3.8V VIN1 = 5V 100 1000
35571 G14
TA = 25C unless otherwise specified Step-Down Switching Regulator 2 1.2V Output Efficiency vs IOUT2
100 90 80 70 60 50 40 30 20 10 0 0.01 0.1 1 10 IOUT2 (mA) VOUT2 = 1.2V VIN2 = 3.8V VIN2 = 5V 100 1000
35571 G15
Step-Down Switching Regulator 1 3.3V Output Efficiency vs IOUT1
Burst Mode OPERATION
PULSE SKIP
PULSE SKIP
VIN = 2.7V
35571 G13
Step-Down Switching Regulator 3 1.8V Output Efficiency vs IOUT3
100 90 80 EFFICIENCY (%) 70 PULSE SKIP IIN (A) 60 50 40 30 20 10 0 0.01 0.1 1 10 IOUT3 (mA) VOUT3 = 1.8V VIN3 = 3.8V VIN3 = 5V 100 1000
35571 G16
Step-Down Switching Regulator Pulse Skip Supply Current vs VINX
400 350 300 250 200 -45C 150 100 2.5 3.0 3.5 4.0 VINX (V) 4.5 5.0
35571 G17
Step-Down Switching Regulator Short-Circuit Current vs Temperature
1200 SHORT-CIRCUIT CURRENT (mA) 1100 1000 900 800 700 600 500 400 -50 -25 0 75 50 25 TEMPERATURE (C) 100 125 400mA BUCK 600mA BUCK
VOUTX = 1.2V IOUTX = 0mA 110C 75C 25C
Burst Mode OPERATION
35571 G18
Step-Down Switching Regulator Output Transient (MODE = 1)
VOUT2 50mV/DIV (AC) VOUT2 50mV/DIV (AC)
Step-Down Switching Regulator Output Transient (MODE = 0)
0.9 0.8 SWITCH IMPEDANCE () 0.7 0.6 0.5 0.4 0.3 0.2 0.1
Step-Down Switching Regulator Switch Impedance vs Temperature
VINX = 3.2V 400mA PMOS
400mA NMOS
VOUT3 50mV/DIV (AC) IOUT2 300mA 5mA VOUT2 = 1.2V VOUT3 = 1.8V IOUT3 = 16mA 50s/DIV
VOUT3 50mV/DIV (AC) IOUT2
35571 G1
600mA PMOS 600mA NMOS
300mA 5mA VOUT2 = 1.2V VOUT3 = 1.8V IOUT3 = 100mA 50s/DIV
35571 G2
0 -50
-25
0 25 50 75 TEMPERATURE (C)
100
125
35571 G21
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LTC3557/LTC3557-1 TYPICAL PERFORMANCE CHARACTERISTICS
600mA Step-Down Switching Regulator Feedback Voltage vs Output Current
0.85 0.84 0.83 0.82 FEEDBACK (V) 0.81 0.80 0.79 0.78 0.77 0.76 0.75 0.1 1 10 100 OUTPUT CURRENT (mA) 3.8V 5V 1000
35571 G22
TA = 25C unless otherwise specified
400mA Step-Down Switching Regulator Feedback Voltage vs Output Current
0.85 0.84 0.83 FEEDBACK (V) Burst Mode OPERATION VOUT2 50mV/DIV(AC)
Step-Down Switching Regulator Start-Up Waveform
Burst Mode OPERATION
0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.75 0.1
VOUT1 1V/DIV 0V IL1 200mA/DIV 0mA EN1 3.8V 5V VOUT2 = 1.2V IOUT2 = 50mA MODE = 1 ROUT1 = 8 100s/DIV
35571 G24
PULSE SKIP
PULSE SKIP
1 10 100 OUTPUT CURRENT (mA)
1000
35571 G23
PIN FUNCTIONS
ILIM0, ILIM1 (Pins 1, 2): Input Current Control Pins. ILIM0 and ILIM1 control the input current limit. See Table 1. Both pins are pulled low by a weak current sink. WALL (Pin 3): Wall Adapter Present Input. Pulling this pin above 4.3V will disconnect the power path from VBUS to VOUT. The ACPR pin will also be pulled low to indicate that a wall adapter has been detected. LDO3V3 (Pin 4): Always On 3.3V LDO Output. The LDO3V3 pin provides a regulated, always-on 3.3V supply voltage. This pin gets its power from VOUT. It may be used for light loads such as a real-time clock or housekeeping microprocessor. A 1F capacitor is required from LDO3V3 to ground if it will be called upon to deliver current. If the LDO3V3 output is not used it should be disabled by connecting it to VOUT. SW1 (Pin 5): Power Transmission (Switch) Pin for Step-Down Switching Regulator 1. V IN1 (Pin 6): Power Input for Step-Down Switching Regulator 1. This pin will generally be connected to VOUT. FB1 (Pin 7): Feedback Input for Step-Down Switching Regulator 1. This pin servos to a fixed voltage of 0.8V when the control loop is complete. MODE (Pin 8): Low Power Mode Enable. When this pin is pulled high, the three step-down switching regulators are set to low power Burst Mode operation. EN1 (Pin 9): Logic Input Enables Step-Down Switching Regulator 1. EN2 (Pin 10): Logic Input Enables Step-Down Switching Regulator 2. EN3 (Pin 11): Logic Input Enables Step-Down Switching Regulator 3. FB3 (Pin 12): Feedback Input for Step-Down Switching Regulator 3. This pin servos to a fixed voltage of 0.8V when the control loop is complete. FB2 (Pin 13): Feedback Input for Step-Down Switching Regulator 2. This pin servos to a fixed voltage of 0.8V when the control loop is complete. RST2 (Pin 14): This is an open-drain output which indicates that step-down switching regulator 2 has settled to its final value. It can be used as a power on reset for the primary microprocessor or to enable the other buck regulators for supply sequencing. SW2 (Pin 15): Power Transmission (Switch) Pin for Step-Down Switching Regulator 2.
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LTC3557/LTC3557-1 PIN FUNCTIONS
VIN2 (Pin 16): Power Input for Step-Down Switching Regulators 2 and 3. This pin will generally be connected to VOUT. SW3 (Pin 17): Power Transmission (Switch) Pin for Step-Down Switching Regulator 3. VNTC (Pin 18): Output Bias Voltage for NTC. A resistor from this pin to the NTC pin will bias the NTC thermistor. NTC (Pin 19): The NTC pin connects to a battery's thermistor to determine if the battery is too hot or too cold to charge. If the battery's temperature is out of range, charging is paused until it drops back into range. A low drift bias resistor is required from VNTC to NTC and a thermistor is required from NTC to ground. If the NTC function is not desired, the NTC pin should be grounded. PROG (Pin 20): Charge Current Program and Charge Current Monitor Pin. Connecting a resistor from PROG to ground programs the charge current: ICHG (A) = 1000V RPROG exceeds the allotted input current from VBUS or if the VBUS power source is removed. VOUT should be bypassed with a low impedance multilayer ceramic capacitor. The total capacitance on VOUT should maintain a minimum of 5F over operating voltage and temperature. VBUS (Pin 24): USB Input Voltage. VBUS will usually be connected to the USB port of a computer or a DC output wall adapter. VBUS should be bypassed with a low impedance multilayer ceramic capacitor. ACPR (Pin 25): Wall Adapter Present Output (Active Low). A low on this pin indicates that the wall adapter input comparator has had its input pulled above its input threshold (typically 4.3V). This pin can be used to drive the gate of an external P-channel MOSFET to provide power to VOUT from a power source other than a USB port. VC (Pin 26): High Voltage Buck Regulator Control Pin. This pin can be used to drive the VC pin of an approved external high voltage buck switching regulator. An external P-channel MOSFET is required to provide power to VOUT with its gate tied to the ACPR pin. The VC pin is designed to work with the LT(R)3480, LT3481 and LT3505. CLPROG (Pin 27): Input Current Program and Input Current Monitor Pin. A resistor from CLPROG to ground determines the upper limit of the current drawn from the VBUS pin (i.e., the input current limit). A precise fraction of the input current, hCLPROG, is sent to the CLPROG pin. The input PowerPath delivers current until the CLPROG pin reaches 2.0V (10x Mode), 1.0V (5x Mode) or 0.2V (1x Mode). Therefore, the current drawn from VBUS will be limited to an amount given by hCLPROG and RCLPROG. In USB applications the resistor RCLPROG should be set to no less than 2.1k. CHRG (Pin 28): Open-Drain Charge Status Output. The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG: charging, not charging (i.e., float charge current less than 1/10th programmed charge current), unresponsive battery (i.e., its voltage remains below 2.8V after 1/2 hour of charging) and battery temperature out of range. CHRG requires a pull-up resistor and/or LED to provide indication. Exposed Pad (Pin 29): Ground. The exposed package pad is ground and must be soldered to the PC board for proper functionality and for maximum heat transfer.
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If sufficient input power is available in constant current mode, this pin servos to 1V. The voltage on this pin always represents the actual charge current. GATE (Pin 21): Ideal Diode Gate Connection. This pin controls the gate of an optional external P-channel MOSFET transistor used to supplement the internal ideal diode. The source of the P-channel MOSFET should be connected to VOUT and the drain should be connected to BAT. It is important to maintain high impedance on this pin and minimize all leakage paths. BAT (Pin 22): Single Cell Li-Ion Battery Pin. Depending on available power and load, a Li-Ion battery on BAT will either deliver system power to VOUT through the ideal diode or be charged from the battery charger. VOUT (Pin 23): Output Voltage of the PowerPath Controller and Input Voltage of the Battery Charger. The majority of the portable product should be powered from VOUT. The LTC3557/LTC3557-1 will partition the available power between the external load on VOUT and the internal battery charger. Priority is given to the external load and any extra power is used to charge the battery. An ideal diode from BAT to VOUT ensures that VOUT is powered even if the load
10
LTC3557/LTC3557-1 BLOCK DIAGRAM
3 WALL WALL DETECT VBUS 25 ACPR 26 VC VC CONTROL VOUT
24
23
18 19
VNTC NTC
BATTERY TEMP MONITOR OSC REF
CHRG 28 CHARGE STATUS EN MODE
ILIM0 1 ILIM1 2 EN1 EN2 EN3 MODE RST2 400mA, 2.25MHz BUCK REGULATOR EN LOGIC EN MODE FB2 ILIM LOGIC OSC 400mA, 2.25MHz BUCK REGULATOR
9 10 11 8 14
OSC
EN MODE GND 29
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+ -
27
CLPROG
INPUT CURRENT LIMIT
CC/CV CHARGER
IDEAL DIODE
+ -
15mV
GATE
21
BAT PROG 3.3V LDO LDO3V3 VIN1
22 20 4 6
600mA, 2.25MHz BUCK REGULATOR
SW1
5
FB1 VIN2
7
16
SW2
15
13
SW3
17
FB3
12
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LTC3557/LTC3557-1 OPERATION
Introduction The LTC3557/LTC3557-1 is a highly integrated power management IC that includes a PowerPath controller, battery charger, an ideal diode, an always-on LDO, three synchronous step-down switching regulators as well as a buck regulator VC controller. Designed specifically for USB applications, the PowerPath controller incorporates a precision input current limit which communicates with the battery charger to ensure that input current never violates the USB specifications. The ideal diode from BAT to VOUT guarantees that ample power is always available to VOUT even if there is insufficient or absent power at VBUS. The LTC3557/LTC3557-1 also has the ability to receive power from a wall adapter or other non-current-limited power source. Such a power supply can be connected to the VOUT pin of the LTC3557/LTC3557-1 through an external device such as a power Schottky or FET as shown in Figure 1.
FROM AC ADAPTER (OR HIGH VOLTAGE BUCK OUTPUT) 26 4.3V (RISING) 3.2V (FALLING) 3 WALL VC OPTIONAL CONTROL FOR HIGH VOLTAGE BUCK REGS LT3480, LT3481 OR LT3505 ACPR 25
The LTC3557/LTC3557-1 has the unique ability to use the output, which is powered by an external supply, to charge the battery while providing power to the load. A comparator on the WALL pin is configured to detect the presence of the wall adapter and shut off the connection to the USB. This prevents reverse conduction from VOUT to VBUS when a wall adapter is present. The LTC3557/LTC3557-1 provides a VC output pin which can be used to drive the VC pin of an external high voltage buck switching regulator such as the LT3480, LT3481, or LT3505 to provide power to the VOUT pin. The VC control circuitry adjusts the regulation point of the switching regulator to a small voltage above the BAT pin voltage. This control method provides a high input voltage, high efficiency battery charger and PowerPath function. An "always on" LDO provides a regulated 3.3V from VOUT. This LDO will be on at all times and can be used to supply up to 25mA.
+ -
FROM USB 24 VBUS
75mV (RISING) 25mV (FALLING) ENABLE VOUT VOUT
BAT
Figure 1. Simplified PowerPath Block Diagram
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+ -
+ - + -
23
USB CURRENT LIMIT IDEAL DIODE CONSTANT CURRENT CONSTANT VOLTAGE BATTERY CHARGER
SYSTEM LOAD
+ -
15mV
GATE
21
OPTIONAL EXTERNAL IDEAL DIODE PMOS
BAT
22
+
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Li-Ion
LTC3557/LTC3557-1 OPERATION
The LTC3557/LTC3557-1 includes three 2.25MHz constant frequency current mode step-down switching regulators providing 400mA, 400mA and 600mA each. All stepdown switching regulators can be programmed for a minimum output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory or other logic circuitry. All step-down switching regulators support 100% duty cycle operation and are capable of operating in Burst Mode operation for highest efficiencies at light loads (Burst Mode operation is pin selectable). No external compensation components are required for the switching regulators. USB PowerPath Controller The input current limit and charge control circuits of the LTC3557/LTC3557-1 are designed to limit input current as well as control battery charge current as a function of IVOUT. VOUT drives the combination of the external load, the three step-down switching regulators, always on 3.3V LDO and the battery charger. If the combined load does not exceed the programmed input current limit, VOUT will be connected to VBUS through an internal 200m P-channel MOSFET. If the combined load at VOUT exceeds the programmed input current limit, the battery charger will reduce its charge current by the amount necessary to enable the external load to be satisfied while maintaining the programmed input current. Even if the battery charge current is set to exceed the allowable USB current, the average input current USB specification will not be violated. Furthermore, load current at VOUT will always be prioritized and only excess available current will be used to charge the battery. The current out of the CLPROG pin is a fraction (1/hCLPROG) of the VBUS current. When a programming resistor is connected from CLPROG to GND, the voltage on CLPROG represents the input current: I VBUS = IBUSQ + VCLPROG *h RCLPROG CLPROG The input current limit is programmed by the ILIM0 and ILIM1 pins. The LTC3557/LTC3557-1 can be configured to limit input current to one of several possible settings as well as be deactivated (USB Suspend). The input current limit will be set by the appropriate servo voltage and the resistor on CLPROG according to the following expression: IVBUS = IBUSQ + IVBUS = IBUSQ + IVBUS = IBUSQ + 0.2V RCLPROG 1V RCLPROG 2V RCLPROG * hCLPROG (1x Mode) * hCLPROG (5x Mode) * hCLPROG (10x Mode)
Under worst-case conditions, the USB specification will not be violated with an RCLPROG resistor of greater than 2.1k. Table 1 shows the available settings for the ILIM0 and ILIM1 pins:
Table 1: Controlled Input Current Limit
ILIM1 0 0 1 1 ILIM0 0 1 0 1 IBUS(LIM) 100mA (1x) 1A (10x) Suspend 500mA (5x)
Notice that when ILIM0 is high and ILIM1 is low, the input current limit is set to a higher current limit for increased charging and current availability at VOUT. This mode is typically used when there is power available from a wall adapter. Ideal Diode from BAT to VOUT The LTC3557/LTC3557-1 has an internal ideal diode as well as a controller for an optional external ideal diode. Both the internal and the external ideal diodes respond quickly whenever VOUT drops below BAT. If the load increases beyond the input current limit, additional current will be pulled from the battery via the ideal diodes. Furthermore, if power to VBUS (USB) or VOUT (external wall power or high voltage regulator) is removed, then all of the application power will be provided
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where IBUSQ and hCLPROG are given in the Electrical Characteristics.
13
LTC3557/LTC3557-1 OPERATION
by the battery via the ideal diodes. The ideal diodes are fast enough to keep VOUT from dropping with just the recommended output capacitor. The ideal diode consists of a precision amplifier that enables an on-chip P-channel MOSFET whenever the voltage at VOUT is approximately 15mV (VFWD) below the voltage at BAT. The resistance of the internal ideal diode is approximately 200m. If this is sufficient for the application, then no external components are necessary. However, if more conductance is needed, an external P-channel MOSFET can be added from BAT to VOUT. The GATE pin of the LTC3557/LTC3557-1 drives the gate of the external P-channel MOSFET for automatic ideal diode control. The source of the MOSFET should be connected to VOUT and the drain should be connected to BAT. Capable of driving a 1nF load, the GATE pin can control an external P-channel MOSFET having extremely low on-resistance. Using the WALL Pin to Detect the Presence of an External Power Source The WALL input pin can be used to identify the presence of an external power source (particularly one that is not subject to a fixed current limit like the USB VBUS input). Typically, such a power supply would be a 5V wall adapter output or the low voltage output of a high voltage buck regulator (specifically, LT3480, LT3481 or LT3505). When the wall adapter output (or buck regulator output) is connected directly to the WALL pin, and the voltage exceeds the WALL pin threshold, the USB power path (from VBUS to VOUT) will be disconnected. Furthermore, the ACPR pin will be pulled low. In order for the presence of an external power supply to be acknowledged, both of the following conditions must be satisfied: 1. The WALL pin voltage must exceed approximately 4.3V. 2. The WALL pin voltage must exceed 75mV above the BAT pin voltage. The input power path (between VBUS and VOUT) is re-enabled and the ACPR pin is pulled high when either of the following conditions is met: 1. The WALL pin voltage falls to within 25mV of the BAT pin voltage. 2. The WALL pin voltage falls below 3.2V. Each of these thresholds is suitably filtered in time to prevent transient glitches on the WALL pin from falsely triggering an event. See the Applications Information section for an explanation of high voltage buck regulator control using the VC pin. Suspend Mode When ILIM0 is pulled low and ILIM1 is pulled high the LTC3557/LTC3557-1 enters Suspend mode to comply with the USB specification. In this mode, the power path between VBUS and VOUT is put in a high impedance state to reduce the VBUS input current to 50A. If no other power source is available to drive WALL and VOUT, the system load connected to VOUT is supplied through the ideal diodes connected to BAT. If an external power source drives WALL and VOUT such that VOUT < VBUS, the Suspend mode VBUS input current can be as high as 200A. 3.3V Always-On Supply The LTC3557/LTC3557-1 includes an ultralow quiescent current low dropout regulator that is always powered. This LDO can be used to provide power to a system pushbutton controller or standby microcontroller. Designed to deliver up to 25mA, the always-on LDO requires a 1F MLCC bypass capacitor for compensation. The LDO is powered from VOUT, and therefore will enter dropout at loads less than 25mA as VOUT falls near 3.3V. If the LDO3V3 output is not used, it should be disabled by connecting it to VOUT. VBUS Undervoltage Lockout (UVLO) An internal undervoltage lockout circuit monitors VBUS and keeps the input current limit circuitry off until VBUS rises above the rising UVLO threshold (3.8V) and at least 50mV above VOUT. Hysteresis on the UVLO turns off the input current limit if VBUS drops below 3.7V or 50mV below VOUT. When this happens, system power at VOUT will be drawn from the battery via the ideal diode. To minimize the possibility of oscillation in and out of UVLO when using resistive input supplies, the input current limit is reduced as VBUS drops below 4.45V typical.
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LTC3557/LTC3557-1 OPERATION
Battery Charger The LTC3557/LTC3557-1 includes a constant current/constant voltage battery charger with automatic recharge, automatic termination by safety timer, low voltage trickle charging, bad cell detection and thermistor sensor input for out of temperature charge pausing. When a battery charge cycle begins, the battery charger first determines if the battery is deeply discharged. If the battery voltage is below VTRKL, typically 2.85V, an automatic trickle charge feature sets the battery charge current to 10% of the programmed value. If the low voltage persists for more than 1/2 hour, the battery charger automatically terminates and indicates via the CHRG pin that the battery was unresponsive. Once the battery voltage is above 2.85V, the battery charger begins charging in full power constant current mode. The current delivered to the battery will try to reach 1000V/ RPROG. Depending on available input power and external load conditions, the battery charger may or may not be able to charge at the full programmed rate. The external load will always be prioritized over the battery charge current. The USB current limit programming will always be observed and only additional current will be available to charge the battery. When system loads are light, battery charge current will be maximized. Charge Termination The battery charger has a built-in safety timer. When the battery voltage approaches the float voltage (4.2V for LTC3557 or 4.1V for LTC3557-1), the charge current begins to decrease as the LTC3557/LTC3557-1 enters constant voltage mode. Once the battery charger detects that it has entered constant voltage mode, the four hour safety timer is started. After the safety timer expires, charging of the battery will terminate and no more current will be delivered. Automatic Recharge After the battery charger terminates, it will remain off drawing only microamperes of current from the battery. If the portable product remains in this state long enough, the battery will eventually self discharge. To ensure that the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below VRECHRG (typically 4.1V for the LTC3557 or 4V for LTC3557-1). In the event that the safety timer is running when the battery voltage falls below VRECHRG, the timer will reset back to zero. To prevent brief excursions below VRECHRG from resetting the safety timer, the battery voltage must be below VRECHRG for more than 1.3ms. The charge cycle and safety timer will also restart if the VBUS UVLO cycles low and then high (e.g., VBUS, is removed and then replaced). Charge Current The charge current is programmed using a single resistor from PROG to ground. 1/1000th of the battery charge current is delivered to PROG which will attempt to servo to 1.000V. Thus, the battery charge current will try to reach 1000 times the current in the PROG pin. The program resistor and the charge current are calculated using the following equations: RPROG = 1000V 1000V , ICHG = ICHG RPROG
In either the constant current or constant voltage charging modes, the PROG pin voltage will be proportional to the actual charge current delivered to the battery. Therefore, the actual charge current can be determined at any time by monitoring the PROG pin voltage and using the following equation: IBAT = VPROG * 1000 RPROG
In many cases, the actual battery charge current, IBAT, will be lower than ICHG due to limited input current available and prioritization with the system load drawn from VOUT. Thermal Regulation To prevent thermal damage to the IC or surrounding components, an internal thermal feedback loop will automatically decrease the programmed charge current if the die temperature rises to approximately 110C. Thermal regulation protects the LTC3557/LTC3557-1 from excessive temperature due to high power operation
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LTC3557/LTC3557-1 OPERATION
or high ambient thermal conditions and allows the user to push the limits of the power handling capability with a given circuit board design without risk of damaging the LTC3557/LTC3557-1 or external components. The benefit of the LTC3557/LTC3557-1 thermal regulation loop is that charge current can be set according to actual conditions rather than worst-case conditions with the assurance that the battery charger will automatically reduce the current in worst-case conditions. Charge Status Indication The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG which include charging, not charging, unresponsive battery and battery temperature out of range. The signal at the CHRG pin can be easily recognized as one of the above four states by either a human or a microprocessor. An open-drain output, the CHRG pin can drive an indicator LED through a current limiting resistor for human interfacing or simply a pull-up resistor for microprocessor interfacing. To make the CHRG pin easily recognized by both humans and microprocessors, the pin is either a DC signal of ON for charging, OFF for not charging, or it is switched at high frequency (35kHz) to indicate the two possible faults, unresponsive battery, and battery temperature out of range. When charging begins, CHRG is pulled low and remains low for the duration of a normal charge cycle. When charging is complete, i.e., the charger enters constant voltage mode and the charge current has dropped to one-tenth of the programmed value, the CHRG pin is released (high impedance). The CHRG pin does not respond to the C/10 threshold if the LTC3557/LTC3557-1 is in input current limit. This prevents false end of charge indications due to insufficient power available to the battery charger. If a fault occurs, the pin is switched at 35kHz. While switching, its duty cycle is modulated between a high and low value at a very low frequency. The low and high duty cycles are disparate enough to make an LED appear to be on or off thus giving the appearance of "blinking". Each of the two faults has its own unique "blink" rate for human recognition as well as two unique duty cycles for machine recognition. Table 2: illustrates the four possible states of the CHRG pin when the battery charger is active.
Table 2: CHRG Output Pin
MODULATION (BLINK) FREQUENCY 0Hz (Lo-Z) 0Hz (Hi-Z) 1.5Hz at 50% 6.1Hz at 50%
STATUS Charging IBAT < C/10 NTC Fault Bad Battery
FREQUENCY 0Hz 0Hz 35kHz 35kHz
DUTY CYCLE 100% 0% 6.25% or 93.75% 12.5% or 87.5%
An NTC fault is represented by a 35kHz pulse train whose duty cycle toggles between 6.25% and 93.75% at a 1.5Hz rate. A human will easily recognize the 1.5Hz rate as a "slow" blinking which indicates the out-of-range battery temperature while a microprocessor will be able to decode either the 6.25% or 93.75% duty cycles as an NTC fault. If a battery is found to be unresponsive to charging (i.e., its voltage remains below VTRKL, typically 2.8V, for 1/2 hour), the CHRG pin gives the battery fault indication. For this fault, a human would easily recognize the frantic 6.1Hz "fast" blink of the LED while a microprocessor would be able to decode either the 12.5% or 87.5% duty cycles as a bad battery fault. Note that the LTC3557/LTC3557-1 is a 3-terminal PowerPath product where system load is always prioritized over battery charging. Due to excessive system load, there may not be sufficient power to charge the battery beyond the trickle charge threshold voltage within the bad battery timeout period. In this case, the battery charger will falsely indicate a bad battery. System software may then reduce the load and reset the battery charger to try again. Although very improbable, it is possible that a duty cycle reading could be taken at the bright-dim transition (low duty cycle to high duty cycle). When this happens the duty cycle reading will be precisely 50%. If the duty cycle reading is 50%, system software should disqualify it and take a new duty cycle reading.
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LTC3557/LTC3557-1 OPERATION
NTC Thermistor The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the battery pack. The NTC circuitry is shown in Figure 8. To use this feature connect the NTC thermistor, RNTC, between the NTC pin and ground and a bias resistor, RNOM, from VNTC to NTC. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC thermistor at 25C (R25). A 100k thermistor is recommended since thermistor current is not measured by the LTC3557/LTC3557-1 and will have to be considered for USB compliance. The LTC3557/LTC3557-1 will pause charging when the resistance of the NTC thermistor drops to 0.54 times the value of R25 or approximately 54k (for a Vishay "Curve 1" thermistor, this corresponds to approximately 40C). If the battery charger is in constant voltage (float) mode, the safety timer also pauses until the thermistor indicates a return to a valid temperature. As the temperature drops, the resistance of the NTC thermistor rises. The LTC3557/ LTC3557-1 is also designed to pause charging when the value of the NTC thermistor increases to 3.25 times the value of R25. For a Vishay "Curve 1" thermistor this resistance, 325k, corresponds to approximately 0C. The hot and cold comparators each have approximately 3C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin disables all NTC functionality. General Purpose Step-Down Switching Regulators The LTC3557/LTC3557-1 includes three 2.25MHz constant frequency current mode step-down switching regulators providing 400mA, 400mA and 600mA each. All step-down switching regulators can be programmed for a minimum output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory or other logic circuitry. All step-down switching regulators support 100% duty cycle operation (low dropout mode) when the input voltage drops very close to the output voltage and are also capable of Burst Mode operation for highest efficiencies at light loads (Burst Mode operation is pin selectable). The step-down switching regulators also include soft-start to limit inrush current when powering on, short-circuit current protection, and switch node slew limiting circuitry to reduce EMI radiation. No external compensation components are required for the switching regulators. A single MODE pin sets all step-down switching regulators in Burst Mode or pulse-skip mode operation, while each regulator is enabled individually through their respective enable pins (EN1, EN2 and EN3). It is recommended that the step-down switching regulator input supplies (VIN1 and VIN2) be connected to the system supply pin (VOUT). This allows the undervoltage lock out circuit on the VOUT pin (VOUT UVLO)to disable the step-down switching regulators when the VOUT voltage drops below VOUT UVLO threshold. If driving the step-down switching regulator input supplies from a voltage other than VOUT the regulators should not be operated outside the specified operating range as operation is not guaranteed beyond this range. Step-Down Switching Regulator Output Voltage Programming Figure 2 shows the step-down switching regulator application circuit. The full-scale output voltage for each step-down switching regulator is programmed using a resistor divider from the step-down switching regulator output connected to the feedback pins (FB1, FB2 and FB3) such that: R1 VOUTx = 0.8V * + 1 R2 Typical values for R1 are in the range of 40k to 1M. The capacitor CFB cancels the pole created by feedback resistors and the input capacitance of the FB pin and also helps to improve transient response for output voltages much greater than 0.8V. A variety of capacitor sizes can be used for CFB but a value of 10pF is recommended for most applications. Experimentation with capacitor sizes between 2pF and 22pF may yield improved transient response.
VIN EN MODE PWM CONTROL MP SWx MN CFB FBx 0.8V GND R1 L VOUTx COUT
R2
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Figure 2. Buck Converter Application Circuit
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LTC3557/LTC3557-1 OPERATION
Step-Down Switching Regulator RST2 Operation The RST2 pin is an open-drain output used to indicate that step-down switching regulator 2 has been enabled and has reached its final voltage. A 230ms delay is included from the time switching regulator 2 reaches 92% of its regulation value to allow a system controller ample time to reset itself. RST2 may be used as a power-on reset to the microprocessor powered by regulator 2 or may be used to enable regulators 1 and/or 3 for supply sequencing. RST2 is an open-drain output and requires a pull-up resistor to the output voltage of regulator 2 or another appropriate power source. Step-Down Switching Regulator Operating Modes The step-down switching regulators include two possible operating modes to meet the noise/power needs of a variety of applications. In pulse-skip mode, an internal latch is set at the start of every cycle, which turns on the main P-channel MOSFET switch. During each cycle, a current comparator compares the peak inductor current to the output of an error amplifier. The output of the current comparator resets the internal latch, which causes the main P-channel MOSFET switch to turn off and the N-channel MOSFET synchronous rectifier to turn on. The N-channel MOSFET synchronous rectifier turns off at the end of the 2.25MHz cycle or if the current through the N-channel MOSFET synchronous rectifier drops to zero. Using this method of operation, the error amplifier adjusts the peak inductor current to deliver the required output power. All necessary compensation is internal to the step-down switching regulator requiring only a single ceramic output capacitor for stability. At light loads in pulse-skip mode, the inductor current may reach zero on each pulse which will turn off the N-channel MOSFET synchronous rectifier. In this case, the switch node (SW1, SW2 or SW3) goes high impedance and the switch node voltage will "ring". This is discontinuous operation, and is normal behavior for a switching regulator. At very light loads in pulse-skip mode, the step-down switching regulators will automatically skip pulses as needed to maintain output regulation. At high duty cycle (VOUTX > VINX/2) it is possible for the inductor current to reverse at light loads causing the stepped down switching regulator to operate continuously. When operating continuously, regulation and low noise output voltage are maintained, but input operating current will increase to a few milliamps. In Burst Mode operation, the step-down switching regulators automatically switch between fixed frequency PWM operation and hysteretic control as a function of the load current. At light loads the step-down switching regulators control the inductor current directly and use a hysteretic control loop to minimize both noise and switching losses. While operating in Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The step-down switching regulator then goes into sleep mode, during which the output capacitor provides the load current. In sleep mode, most of the switching regulator's circuitry is powered down, helping conserve battery power. When the output voltage drops below a pre-determined value, the step-down switching regulator circuitry is powered on and another burst cycle begins. The sleep time decreases as the load current increases. Beyond a certain load current point (about 1/4 rated output load current) the step-down switching regulators will switch to a low noise constant frequency PWM mode of operation, much the same as pulse-skip operation at high loads. For applications that can tolerate some output ripple at low output currents, Burst Mode operation provides better efficiency than pulse-skip at light loads. The step-down switching regulators allow mode transition on-the-fly, providing seamless transition between modes even under load. This allows the user to switch back and forth between modes to reduce output ripple or increase low current efficiency as needed. Burst Mode operation is set by driving the MODE pin high, while pulse-skip mode is achieved by driving the MODE pin low. Step-Down Switching Regulator in Shutdown The step-down switching regulators are in shutdown when not enabled for operation. In shutdown all circuitry in the step-down switching regulator is disconnected from the switching regulator input supply leaving only a few nanoamps of leakage current. The step-down switching regulator outputs are individually pulled to ground through a 10k resistor on the switch pin (SW1, SW2 or SW3) when in shutdown.
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LTC3557/LTC3557-1 OPERATION
Step-down Switching Regulator Dropout Operation It is possible for a step-down switching regulator's input voltage to approach its programmed output voltage (e.g., a battery voltage of 3.4V with a programmed output voltage of 3.3V). When this happens, the PMOS switch duty cycle increases until it is turned on continuously at 100%. In this dropout condition, the respective output voltage equals the regulator's input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. Step-Down Switching Regulator Soft-Start Operation Soft-start is accomplished by gradually increasing the peak inductor current for each step-down switching regulator over a 500s period. This allows each output to rise slowly, helping minimize inrush current required to charge up the switching regulator output capacitor. A soft-start cycle occurs whenever a given switching regulator is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless output transition when actively changing between operating modes. Step-Down Switching Regulator Switching Slew Rate Control The step-down switching regulators contain new patentpending circuitry to limit the slew rate of the switch node (SW1, SW2 and SW3). This new circuitry is designed to transition the switch node over a period of a couple nanoseconds, significantly reducing radiated EMI and conducted supply noise while maintaining high efficiency. Step-Down Switching Regulator Low Supply Operation An undervoltage lockout (UVLO) circuit on VOUT shuts down the step-down switching regulators when VOUT drops below about 2.7V. It is recommended that the step-down switching regulators input supplies be connected to the power path output (VOUT). This UVLO prevents the step-down switching regulators' from operating at low supply voltages where loss of regulation or other undesirable operation may occur. If driving the step-down switching regulator input supplies from a voltage other than the VOUT pin, the regulators should not be operate outside the specified operating range as operation is not guaranteed beyond this range. Step-Down Switching Regulator Inductor Selection Many different sizes and shapes of inductors are available from numerous manufacturers. Choosing the right inductor from such a large selection of devices can be overwhelming, but following a few basic guidelines will make the selection process much simpler. The step-down converters are designed to work with inductors in the range of 2.2H to 10H. For most applications a 4.7H inductor is suggested for step-down switching regulators providing up to 400mA of output current while a 3.3H inductor is suggested for step-down switching regulators providing up to 600mA. Larger value inductors reduce ripple current, which improves output ripple voltage. Lower value inductors result in higher ripple current and improved transient response time, but will reduce the available output current. To maximize efficiency, choose an inductor with a low DC resistance. For a 1.2V output, efficiency is reduced about 2% for 100m series resistance at 400mA load current, and about 2% for 300m series resistance at 100mA load current. Choose an inductor with a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short circuit is a possible condition, the inductor should be rated to handle the maximum peak current specified for the step-down converters. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or Permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher core and DCR losses, and will not give the best efficiency. The choice of which style inductor to use often depends more on the price vs size, performance, and any radiated EMI requirements than on what the step-down switching regulators requires to operate.
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LTC3557/LTC3557-1 OPERATION
The inductor value also has an effect on Burst Mode operation. Lower inductor values will cause Burst Mode switching frequency to increase. Table 3 shows several inductors that work well with the step-down switching regulators. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors. Step-Down Switching Regulator Input/Output Capacitor Selection Low ESR (equivalent series resistance) ceramic capacitors should be used at both step-down switching regulator outputs as well as at each step-down switching regulator input supply. Only X5R or X7R ceramic capacitors should be used because they retain their capacitance over wider voltage and temperature ranges than other ceramic types. A 10F output capacitor is sufficient for the step-down switching regulator outputs. For good transient response and stability the output capacitor for step-down switching regulators should retain at least 4F of capacitance over operating temperature and bias voltage. Each switching regulator input supply should be bypassed with a 2.2F capacitor. Consult with capacitor manufacturers for detailed information on their selection and specifications of ceramic capacitors. Many manufacturers now offer very thin (<1mm tall) ceramic capacitors ideal for use in height-restricted designs. Table 4 shows a list of several ceramic capacitor manufacturers.
Table 4. Ceramic Capacitor Manufacturers
AVX Murata Taiyo Yuden Vishay Siliconix TDK www.avxcorp.com www.murata.com www.t-yuden.com www.vishay.com www.tdk.com
Table 3. Recommended Inductors for Step-Down Switching Regulators
INDUCTOR TYPE DE2818C D312C DE2812C CDRH3D16 CDRH2D11 CLS4D09 SD3118 SD3112 SD12 SD10 LPS3015 *Typical DCR L (H) 4.7 3.3 4.7 3.3 4.7 3.3 4.7 3.3 4.7 3.3 4.7 4.7 3.3 4.7 3.3 4.7 3.3 4.7 3.3 4.7 3.3 MAX IDC (A) 1.25 1.45 0.79 0.90 1.2 1.4 0.9 1.1 0.5 0.6 0.75 1.3 1.59 0.8 0.97 1.29 1.42 1.08 1.31 1.1 1.3 MAX DCR () 0.072 0.053 0.24 0.20 0.13* 0.105* 0.11 0.085 0.17 0.123 0.19 0.162 0.113 0.246 0.165 0.117* 0.104* 0.153* 0.108* 0.2 0.13 SIZE in mm (L x W x H) 3.0 x 2.8 x 1.8 3.0 x 2.8 x 1.8 3.6 x 3.6 x 1.2 3.6 x 3.6 x 1.2 3.0 x 2.8 x 1.2 3.0 x 2.8 x 1.2 4.0 x 4.0 x 1.8 4.0 x 4.0 x 1.8 3.2 x 3.2 x 1.2 3.2 x 3.2 x 1.2 4.9 x 4.9 x 1.0 3.1 x 3.1 x 1.8 3.1 x 3.1 x 1.8 3.1 x 3.1 x 1.2 3.1 x 3.1 x 1.2 5.2 x 5.2 x 1.2 5.2 x 5.2 x 1.2 5.2 x 5.2 x 1.0 5.2 x 5.2 x 1.0 3.0 x 3.0 x 1.5 3.0 x 3.0 x 1.5 Coil Craft www.coilcraft.com Cooper www.cooperet.com Sumida www.sumida.com MANUFACTURER Toko www.toko.com
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LTC3557/LTC3557-1 APPLICATIONS INFORMATION
External HV Buck Control Through the VC Pin The WALL, ACPR and VC pins can be used in conjunction with an external high voltage buck regulator such as the LT(R)3480, LT3481 or LT3505 to provide power directly to the VOUT pin through a power P-channel MOSFET as shown in Figures 3-5 (consult the factory for a complete list of approved high voltage buck regulators). When the WALL pin voltage exceeds 4.3V, VC pin control circuitry is enabled and drives the VC pin of the LT3480, LT3481 or LT3505. The VC pin control circuitry is designed so that no
HVIN 8V TO 38V (TRANSIENTS TO 60V) 4 VIN BOOST 150k LT3480 5 RUN/SS SW 10 RT 40.2k 7 NC BD 6 NC FB GND VC 11 9 Si2333DS UP TO 2A VOUT COUT Si2333DS (OPT) 2 0.47F 3 6.8H 499k 22F 100k
compensation components are required on the VC node. The voltage at the VOUT pin is regulated to the larger of (BAT + 300mV) or 3.6V as shown in Figures 6 and 7. The feedback network of the high voltage regulator should be set to generate an output voltage higher than 4.4V (be sure to include the output voltage tolerance of the buck regulator). The VC control of the LTC3557 overdrives the local VC control of the external high voltage buck. Therefore, once the VC control is enabled, the output voltage is set independent of the buck regulator feedback network.
4.7F
68nF
DFLS240L 1 8
LT3480 HIGH VOLTAGE BUCK CIRCUITRY
26
3
25
VC WALL ACPR 23 VOUT 21 LTC3557 LTC3557-1 GATE 22 BAT
+
BAT Li-Ion
35571 F03
Figure 3. LT3480 Buck Control Using VC (800kHz Switching)
4 4.7F 68nF 2 0.47F 6.8H 549k 22F 200k
HVIN 8V TO 34V
VIN
BOOST
150k LT3481 5 3 RUN/SS SW 10 7 BIAS RT 60.4k 1 BD 6 8 NC FB GND VC 11 9
DFLS240L
LT3481 HIGH VOLTAGE BUCK CIRCUITRY
Si2333DS
UP TO 2A VOUT COUT Si2333DS (OPT)
26
3
25
VC WALL ACPR 23 VOUT 21 LTC3557 LTC3557-1 GATE 22 BAT
+
BAT Li-Ion
35571 F04
Figure 4. LT3481 Buck Control Using VC (800kHz Switching)
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21
LTC3557/LTC3557-1 APPLICATIONS INFORMATION
HVIN 8V TO 36V 1F 3 68nF 150k 4 BZT52C16T 20k 806k 6 RT GND LT3505 HIGH VOLTAGE BUCK CIRCUITRY 5, 9 FB VC 8 Si2333DS UP TO 1.2A VOUT COUT Si2333DS (OPT) 7 VIN BOOST 1 0.1F 1N4148 6.8H 49.9k 10F 10.0k LT3505 2 SW SHDN
MBRM140
26
3
25
VC WALL ACPR 23 VOUT 21 LTC3557 LTC3557-1 GATE 22 BAT
+
BAT Li-Ion
35571 F05
Figure 5. LT3505 Buck Control Using VC (2.2MHz Switching with Frequency Foldback)
5.0
4.5
VOUT (V)
4.0
This technique provides a significant efficiency advantage over the use of a 5V buck to drive the battery charger. With a simple 5V buck output driving VOUT, battery charger efficiency is approximately: CHARGER = BUCK *
IO = 0.0A IO = 0.75A IO = 1.5A BAT 3 3.5 BAT (V) 4 4.5
35571 F06
3.5
VBAT 5V
3.0
2.5 2.5
Figure 6. LTC3557 VOUT Voltage vs Battery Voltage with the LT3480
5.0
where BUCK is the efficiency of the high voltage buck regulator and 5V is the output voltage of the buck regulator. With a typical buck efficiency of 87% and a typical battery voltage of 3.8V, the total battery charger efficiency is approximately 66%. Assuming a 1A charge current, this works out to nearly 2W of power dissipation just to charge the battery! With the VC control technique, battery charger efficiency is approximately: CHARGER = BUCK * VBAT 0.3V + VBAT
4.5
VOUT (V)
4.0
3.5
3.0
IO = 0.0A IO = 0.6A BAT 3 3.5 BAT (V) 4 4.5
35571 F07
2.5 2.5
Figure 7. LTC3557-1VOUT Voltage vs Battery Voltage with the LT3505
With the same assumptions as above, the total battery charger efficiency is approximately 81%. This example works out to just 900mW of power dissipation. For applications, component selection and board layout information beyond those listed here please refer to the respective LT3480, LT3481 or LT3505 data sheet.
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22
LTC3557/LTC3557-1 APPLICATIONS INFORMATION
Alternate NTC Thermistors and Biasing The LTC3557/LTC3557-1 provides temperature qualified charging if a grounded thermistor and a bias resistor are connected to NTC. By using a bias resistor whose value is equal to the room temperature resistance of the thermistor (R25) the upper and lower temperatures are pre-programmed to approximately 40C and 0C, respectively (assuming a Vishay "Curve 1" thermistor). The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value or by adding a second adjustment resistor to the circuit. If only the bias resistor is adjusted, then either the upper or the lower threshold can be modified but not both. The other trip point will be determined by the characteristics of the thermistor. Using the bias resistor in addition to an adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with the constraint that the difference between the upper and lower temperature thresholds cannot decrease. Examples of each technique are given below. NTC thermistors have temperature characteristics which are indicated on resistance-temperature conversion tables. The Vishay-Dale thermistor NTHS0603N011-N1003F used , in the following examples, has a nominal value of 100k and follows the Vishay "Curve 1" resistance-temperature characteristic. In the explanation below, the following notation is used. R25 = Value of the Thermistor at 25C RNTC|COLD = Value of thermistor at the cold trip point RNTC|HOT = Value of the thermistor at the hot trip point rCOLD = Ratio of RNTC|COLD to R25 rHOT= Ratio of RNTC|HOT to R25 RNOM = Primary thermistor bias resistor (see Figure 8) R1 = Optional temperature range adjustment resistor (see Figure 9) The trip points for the LTC3557/LTC3557-1's temperature qualification are internally programmed at 0.349 * VVNTC for the hot threshold and 0.765 * VVNTC for the cold threshold. Therefore, the hot trip point is set when: RNTC|HOT RNOM + RNTC|HOT * VVNTC = 0.349 * VVNTC
and the cold trip point is set when: RNTC|COLD RNOM + RNTC|COLD * VVNTC = 0.765 * VVNTC
Solving these equations for RNTC|COLD and RNTC|HOT results in the following: RNTC|HOT = 0.536 * RNOM and RNTC|COLD = 3.25 * RNOM By setting RNOM equal to R25, the above equations result in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios to the Vishay Resistance-Temperature Curve 1 chart gives a hot trip point of about 40C and a cold trip point of about 0C. The difference between the hot and cold trip points is approximately 40C. By using a bias resistor, RNOM, different in value from R25, the hot and cold trip points can be moved in either direction. The temperature span will change somewhat due to the non-linear behavior of the thermistor. The following equations can be used to easily calculate a new value for the bias resistor: RNOM = RNOM = rHOT * R25 0.536 rCOLD * R25 3.25
where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations are linked. Therefore, only one of the two trip points can be chosen, the other is determined by the default ratios designed in the IC. Consider an example where a 60C hot trip point is desired.
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LTC3557/LTC3557-1 APPLICATIONS INFORMATION
From the Vishay Curve 1 R-T characteristics, rHOT is 0.2488 at 60C. Using the above equation, RNOM should be set to 46.4k. With this value of RNOM, the cold trip point is about 16C. Notice that the span is now 44C rather than the previous 40C. This is due to the decrease in "temperature gain" of the thermistor as absolute temperature increases. The upper and lower temperature trip points can be independently programmed by using an additional bias resistor as shown in Figure 9. The following formulas can be used to compute the values of RNOM and R1: RNOM = rCOLD - rHOT * R25 2.714 the nearest 1% value is 12.7k. The final solution is shown in Figure 9 and results in an upper trip point of 45C and a lower trip point of 0C. Battery Charger Stability Considerations The LTC3557/LTC3557-1's battery charger contains both a constant voltage and a constant current control loop. The constant voltage loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1F from BAT to GND. Furthermore, a 4.7F capacitor in series with a 0.2 to 1 resistor from BAT to GND is required to keep ripple voltage low when the battery is disconnected. High value, low ESR multilayer ceramic chip capacitors reduce the constant voltage loop phase margin, possibly resulting in instability. Ceramic capacitors up to 22F may be used in parallel with a battery, but larger ceramics should be decoupled with 0.2 to 1 of series resistance. In constant current mode, the PROG pin is in the feedback loop rather than the battery voltage. Because of the additional pole created by any PROG pin capacitance, capacitance on this pin must be kept to a minimum. With
VNTC 18 RNOM 105k NTC 19 0.765 * VVNTC
R1= 0.536 * RNOM - rHOT * R25 For example, to set the trip points to 0C and 45C with a Vishay Curve 1 thermistor choose RNOM = 3.266 - 0.4368 * 100k = 104.2k 2.714
the nearest 1% value is 105k. R1 = 0.536 * 105k - 0.4368 * 100k = 12.6k
VNTC 18 RNOM 100k NTC 19 0.765 * VVNTC
NTC BLOCK
NTC BLOCK
-
TOO_COLD
-
TOO_COLD
+
+
RNTC 100k 0.349 * VVNTC
-
TOO_HOT
R1 12.7k 0.349 * VVNTC RNTC 100k
-
TOO_HOT
+
+
+
NTC_ENABLE 0.017 * VVNTC
+
NTC_ENABLE 0.017 * VVNTC
-
35571 F08
-
35571 F09
Figure 8. Typical NTC Thermistor Circuit
Figure 9. NTC Thermistor Circuit with Additional Bias Resistor
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LTC3557/LTC3557-1 APPLICATIONS INFORMATION
no additional capacitance on the PROG pin, the battery charger is stable with program resistor values as high as 25k. However, additional capacitance on this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept above 100kHz. Therefore, if the PROG pin has a parasitic capacitance, CPROG, the following equation should be used to calculate the maximum resistance value for RPROG: RPROG 1 2 * 100kHz * CPROG It is not necessary to perform any worst-case power dissipation scenarios because the LTC3557/LTC3557-1 will automatically reduce the charge current to maintain the die temperature at approximately 110C. However, the approximate ambient temperature at which the thermal feedback begins to protect the IC is: TA = 110C - PD * JA Example: Consider the LTC3557/LTC3557-1 operating from a wall adapter with 5V (VOUT) providing 1A (IBAT) to charge a Li-Ion battery at 3.3V (BAT). Also assume PD(SW1) = PD(SW2) = PD(SW3) = 0.05W, so the total power dissipation is: PD = (5V - 3.3V) * 1A + 0.15W = 1.85W The ambient temperature above which the LTC3557/ LTC3557-1 will begin to reduce the 1A charge current, is approximately: TA = 110C - 1.85W * 37C/W = 42C The LTC3557/LTC3557-1 can be used above 42C, but the charge current will be reduced below 1A. The charge current at a given ambient temperature can be approximated by: PD = 110C - TA JA
Printed Circuit Board Power Dissipation Considerations In order to be able to deliver maximum charge current under all conditions, it is critical that the Exposed Pad on the backside of the LTC3557/LTC3557-1 package is soldered to a ground plane on the board. Correctly soldered to a 2500mm2 ground plane on a double-sided 1oz copper board, the LTC3557/LTC3557-1 has a thermal resistance (JA) of approximately 37C/W. Failure to make good thermal contact between the Exposed Pad on the backside of the package and an adequately sized ground plane will result in thermal resistances far greater than 37C/W. The conditions that cause the LTC3557/LTC3557-1 to reduce charge current due to the thermal protection feedback can be approximated by considering the power dissipated in the part. For high charge currents and a wall adapter applied to VOUT, the LTC3557/LTC3557-1 power dissipation is approximately: PD = (VOUT - BAT) * IBAT + PD(SW1) + PD(SW2) + PD(SW3) where, PD is the total power dissipated, VOUT is the supply voltage, BAT is the battery voltage and IBAT is the battery charge current. PD(SWx) is the power loss by the step-down switching regulators. The power loss for a step-down switching regulator can be calculated as follows: PD(SWx) = (OUTx * IOUT) * (100 - Eff)/100 where OUTx is the programmed output voltage, IOUT is the load current and Eff is the % efficiency which can be measured or looked up on an efficiency graph for the programmed output voltage.
= (VOUT - BAT) * IBAT + PD(SW1) + PD(SW2) + PD(SW3) thus: 110C - TA - PD(SW1) - PD(SW2) - PD(SW3) JA IBAT = VOUT - BAT Consider the above example with an ambient temperature of 55C. The charge current will be reduced to approximately: 110C - 55C - 0.15W 37C/W IBAT = 5V - 3.3V = 1.49W - 0.15W = 786mA 1.7V
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LTC3557/LTC3557-1 APPLICATIONS INFORMATION
If an external buck switching regulator controlled by the LTC3557/LTC3557-1 VC pin is used instead of a 5V wall adapter we see a significant reduction in power dissipated by the LTC3557/LTC3557-1. This is because the external buck switching regulator will drive the PowerPath output (VOUT) to about 3.6V with the battery at 3.3V. If you go through the example above and substitute 3.6V for VOUT we see that thermal regulation does not kick in until about 93C. Thus, the external regulator not only allows higher charging currents, but lower power dissipation means a cooler running application. Printed Circuit Board Layout Considerations When laying out the printed circuit board, the following list should be followed to ensure proper operation of the LTC3557/LTC3557-1: 1. The Exposed Pad of the package (Pin 29) should connect directly to a large ground plane to minimize thermal and electrical impedance. 2. The trace connecting the step-down switching regulator input supply pins (VIN1 and VIN2) and their respective decoupling capacitors should be kept as short as possible. The GND side of these capacitors should connect directly to the ground plane of the part. These capacitors provide the AC current to the internal power MOSFETs and their drivers. It's important to minimize inductance from these capacitors to the pins of the LTC3557/LTC3557-1. Connect VIN1 and VIN2 to VOUT through a short low impedance trace. 3. The switching power traces connecting SW1, SW2 and SW3 to their respective inductors should be minimized to reduce radiated EMI and parasitic coupling. Due to the large voltage swing of the switching nodes, sensitive nodes such as the feedback nodes (FB1, FB2 and FB3) should be kept far away or shielded from the switching nodes or poor performance could result. 4. Connections between the step-down switching regulator inductors and their respective output capacitors should be kept as short as possible. The GND side of the output capacitors should connect directly to the thermal ground plane of the part. 5. Keep the feedback pin traces (FB1, FB2 and FB3) as short as possible. Minimize any parasitic capacitance between the feedback traces and any switching node (i.e., SW1, SW2, SW3 and logic signals). If necessary shield the feedback nodes with a GND trace 6) Connections between the LTC3557/LTC3557-1 power path pins (VBUS and VOUT) and their respective decoupling capacitors should be kept as short as possible. The GND side of these capacitors should connect directly to the ground plane of the part. VOUT should be decoupled with a 10F or greater ceramic capacitor as close as possible to the LTC3557/LTC3557-1.
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LTC3557/LTC3557-1 TYPICAL APPLICATION
HVIN 8V TO 38V (TRANSIENTS TO 60V) 4 4.7F 68nF 150k 5 VIN RUN/SS RT SYNC PG GND 11 BD FB VC 9 Si2333DS 1 8 100k BOOST LT3480 SW 2 0.47F 3 6.8H 499k 22F OPTIONAL HIGH VOLTAGE BUCK INPUT
10 40.2k NC NC 6 7
DFLS240L
26 USB OR 5V WALL ADAPTER 24 10F
3
25 10F 23 2.2F 6 2.2F 16 28 21 22 4 3.3V 25mA 1F ALWAYS ON Si2333DS (OPT) 510
VOUT
VC WALL ACPR VBUS VOUT BVIN1
2.1k 27 2k 20
CLPROG BVIN2 PROG CHRG GATE VNTC NTC BAT LDO3V3
100k 18 100k NTC 19
+
BAT Li-Ion
LTC3557/ LTC3557-1 1 2 9 PMIC CONTROL 10 11 8 ILIM0 ILIM1 EN1 EN2 EN3 MODE FB3 RST2 SW2 FB2 SW3 FB1 SW1 5 7 17 12 14 15 13 4.7H 10pF 100k 232k 4.7H 10pF 806k 649k 3.3H 10pF 1.02M VOUT1 3.3V 10F 600mA VOUT3 1.8V 10F 400mA
324k
RST2 VOUT2 1.2V 10F 400mA
35571 TA02
464k
GND 29
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3557/LTC3557-1 PACKAGE DESCRIPTION
UF Package 28-Lead Plastic QFN (4mm x 4mm)
(Reference LTC DWG # 05-08-1721 Rev A)
BOTTOM VIEW--EXPOSED PAD 4.00 0.10 (4 SIDES) 0.70 0.05 PIN 1 TOP MARK (NOTE 6) 0.75 0.05 R = 0.05 TYP R = 0.115 TYP PIN 1 NOTCH R = 0.20 TYP OR 0.35 45 CHAMFER
27 28 0.40 1 2 0.05
2.64 0.10 (4-SIDES)
0.20 0.40 BSC
PACKAGE OUTLINE 0.05
(UF28) QFN 0106 REVA
0.200 REF 0.00 - 0.05
0.20
0.05
0.40 BSC
DED SOLDER PAD PITCH AND DIMENSIONS R MASK TO AREAS THAT ARE NOT SOLDERED S NOT A JEDEC PACKAGE OUTLINE NOT TO SCALE SIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
RELATED PARTS
PART NUMBER Power Management LTC3455 LTC3456 LTC3555 Dual DC/DC Converter with USB Power Management and Efficiency >96%, Accurate USB Current Limiting (500mA/100mA), Li-Ion Battery Charger 4mm x 4mm 24-Pin QFN Package 2-Cell Multi-Output DC/DC Converter with USB Power Manager Switching USB Power Manager with Li-Ion/Polymer Charger, Triple Synchronous Buck Converter + LDO Seamless Transition Between 2-Cell Battery, USB and AC Wall Adapter Input Power Sources, QFN Package Complete Multifunction PMIC: Switch Mode Power Manager and Three Buck Regulators + LDO, Charge Current Programmable up to 1.5A from Wall Adapter Input, Thermal Regulation Synchronous Buck Converters Efficiency: >95%, ADJ Outputs: 0.8V to 3.6V at 400mA/400mA/1A Bat-Track Adaptive Output Control, 200m Ideal Diode, 4mm x 5mm 28-Pin QFN Package Adjustable Synchronous Buck Converters, Efficiency: >90%, Outputs: Down to 0.8V at 400mA for each, Charge Current Programmable up to 950mA, USB Compatible, 3mm x 3mm 16-Pin QFN Package Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 200m Ideal Diode, 4mm x 4mm 16-Pin QFN Package Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 50m Ideal Diode, 4mm x 4mm 24-Pin QFN Package Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 200m Ideal Diode with <50m Option, 4mm x 3mm 14-Pin DFN Package DESCRIPTION COMMENTS
LTC3559
Linear USB Li-Ion/Polymer Battery Charger with Dual Synchronous Buck Converter
Battery Chargers LTC4055 LTC4066 LTC4085 USB Power Controller and Battery Charger USB Power Controller and Li-Ion Battery Charger with Low Loss Ideal Diode USB Power Manager with Ideal Diode Controller and Li-Ion Charger
LTC4088
High Efficiency USB Power Manager and Battery Charger Maximizes Available Power from USB Port, Bat-Track, "Instant On" Operation, 1.5A Max Charge Current, 180m Ideal Diode with <50m Option, 3.3V/25mA Always On LDO, 4mm x 3mm 14-Pin DFN Package USB Power Manager with Ideal Diode Controller and High Efficiency Li-Ion Battery Charger 1.2A Charger, 6V to 36V (40VMAX), 200m Ideal Diode with 50m Option, 6mm x 3mm 22-Pin DFN Package
35571fc
LTC4089/LTC4089-5
28 Linear Technology Corporation
(408) 432-1900
LT 0808 REV C * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
FAX: (408) 434-0507 www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2007

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